Power-supply device

ABSTRACT

A step-down type DC—DC power-supply device implements both the stabilization of the control loop and the responsibility at the same time. In the power-supply device, an output power signal is fed back to an error amplifier after having passed through a CR smoothing filter provided independently of a power LC smoothing filter. Also, independently of the duty controls over Power MOSFETs, i.e., upper-side/lower-side semiconductor switching components in the steady state, an output from the power LC smoothing filter is added to an upper and lower limit-mode-equipped control circuit, thereby, at the transient state, forcefully setting the duty α at either 0% or 100%.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a power-supply device where,independently of a power LC smoothing filter, a signal is caused to passthrough a CR smoothing filter and is then fed back so that the controlloop will be stabilized.

2. Description of the Related Art

A prior art on the loop stabilizing method for a power-supply device hasbeen described in “Low-Voltage On-Board DC/DC Modules for NextGenerations of Data Processing Circuits”, Zhang et al., IEEE Tran. onPower Elect. Vol. 11, No. 2, March 1996. In the power-supply deviceaccording to the prior art, a signal is fed back to an error amplifierfrom a power LC smoothing filter. Then, the error amplifier compensatesthe phase, thereby implementing the stabilization of the control loop.In this prior art, an aluminum electrolytic capacitor is used as thepower LC smoothing filter.

U.S. Pat. No. 5,877,611 discloses a power supply system in which anoutput of a CR smoothing filter connected across an inductor of anoutput LC smoothing filter is fed back to an error amplifier having alow input impedance. In the U.S. patent prior art, voltage and currentsignals of a power supply output are extracted using the CR smoothingfilter, so that the resistance value of the CR smoothing filter must beset to be small. The component constants of the CR smoothing filter area capacitance of 0.47 μF and a resistance of 100 Ω. Accordingly, the CRsmoothing filter having such constants cannot be formed on chip in apower supply IC and must be formed externally of the IC chip, resultingin a problem that the power supply device cannot be made in small sizetotally.

SUMMARY OF THE INVENTION

In order to downsize the power-supply device, instead of using thealuminum electrolytic capacitor as the power LC smoothing filter, therehas occurred a necessity for using a ceramic capacitor of a chip-part asthe power LC smoothing filter. However, the equivalent series resistance(ESR) of the chip ceramic capacitor is equal to several mΩ, which isconsiderably small. What is more, the ceramic capacitors are connectedin parallel under an actual use condition. Accordingly, the total of theESRs in this case becomes less than 1 mΩ, which is even smaller. Thismakes it impossible to expect the damping of the ESR as is expected inthe case of using the aluminum electrolytic capacitor. Consequently, itbecomes difficult to stabilize the control loop.

In the above-described prior art, when using the ceramic capacitor withthe small ESR as the power LC smoothing filter, it becomes impossible toexpect the damping effect of the ESR. This causes a signal to oscillate,thereby making the phase compensation difficult. Also, if, in the priorart, it were to become possible to implement the phase compensation bynarrowing the operation bandwidth of the error amplifier, a responsefrom the power-supply is delayed exceedingly. Moreover, in modifying theLC smoothing filter's constants, there exists a troublesome task ofadjusting the phase compensation condition of the error amplifier oneach that occasion.

It is an object of the present invention to provide a power-supplydevice that employs a novel control method where, independently of apower LC smoothing filter, a signal is caused to pass through a CRsmoothing filter and is then fed back so that the control loop will bestabilized.

A power-supply device according to one aspect of the present inventionis as follows: In the control loop of the power-supply device of astep-down type DC—DC converter, a CR smoothing filter is providedindependently of a power LC smoothing filter. Moreover, a signalcorresponding to the output power is fed back to an error amplifierafter having passed through the CR smoothing filter.

Also, a power-supply device according to another aspect of the presentinvention includes the following unit: Independently of the dutycontrols over Power MOSFETs, i.e., upper-side/lower-side semiconductorswitching components in the steady state, the unit adds the output froma power LC smoothing filter to an upper and lower limit value detectingcircuit, thereby, at the transient state, forcefully setting the duty ateither 0% or 100%.

Moreover, a power-supply device according to still another aspect of thepresent invention is as follows: The power-supply device includespower-supply device units prepared in plural number. In order to performa parallel operation of these power-supply device units, thepower-supply device further includes an oscillator and a phase shiftcircuit that the plural power-supply device units have in common.Furthermore, in the steady state, phases of driving pulses ofupper-side/lower-side Power MOSFETs in the respective power-supplydevice units are respectively shifted to phases that result fromdividing 360° by the number of the parallelism. At the transient state,all of the parallel power-supply device units are operated by drivingpulses of one and the same phase.

Other objects, features and advantages of the invention will becomeapparent from the following description of the embodiments of theinvention taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit block diagram for illustrating a power-source deviceof a first embodiment in the present invention;

FIG. 2 is an explanatory diagram for explaining an IC where a CR filteris built in a semiconductor chip in the power-supply device in FIG. 1;

FIG. 3 is a circuit block diagram for illustrating a power-supply deviceof a second embodiment in the present invention;

FIG. 4 is an explanatory diagram for explaining an IC where a CR filteris built in a semiconductor chip in the power-supply device in FIG. 3;

FIG. 5 is a circuit block diagram for illustrating a power-supply deviceof a third embodiment in the present invention;

FIG. 6 is a circuit diagram for illustrating the details in FIG. 5;

FIG. 7 is a diagram for illustrating the operation state mode in FIG. 6;

FIG. 8 is a circuit block diagram for illustrating a power-supply deviceof a fourth embodiment in the present invention;

FIG. 9 is a circuit block diagram for illustrating another power-supplydevice of the fourth embodiment;

FIG. 10 is a circuit block diagram for illustrating still anotherpower-supply device of the fourth embodiment;

FIG. 11 is a circuit block diagram for illustrating a multi-phasepower-source device of a fifth embodiment in the present invention;

FIG. 12 is a circuit diagram for illustrating the details in FIG. 11;

FIG. 13 is a diagram for illustrating the operation state mode in FIG.12;

FIG. 14 is a circuit block diagram for illustrating an example of thechip configuration of a power-source device of a sixth embodiment in thepresent invention;

FIG. 15 is an explanatory diagram for explaining a VID code input D/Aconverter applied to FIG. 14;

FIG. 16 is a circuit block diagram for illustrating a multi-phasecompatible chip of a seventh embodiment in the present invention;

FIG. 17 is an explanatory diagram for explaining the printed wiringboard implementation of a power-source control IC of an eighthembodiment;

FIG. 18 is an explanatory diagram for explaining a HDD device of a ninthembodiment;

FIG. 19 is an explanatory diagram for explaining a tenth embodiment inthe present invention;

FIG. 20 is an explanatory diagram for illustrating another embodiment ofa pulse-width modulation oscillator PWM;

FIG. 21 is an explanatory diagram for explaining an eleventh embodimentin the present invention applied to a commercially-availablepower-source IC; and

FIG. 22 is a diagram for illustrating the operation state mode in FIG.21.

DETAILED DESCRIPTION OF THE EMBODIMENTS

Referring to the accompanying drawings, the explanation will be givenbelow concerning the details of the present invention.

Embodiment 1

FIG. 1 illustrates a power-supply device of the present embodiment. InFIG. 1, reference notations Vi and Vo denote an input terminal and anoutput terminal, respectively. An upper-side Power MOSFET Q1 isconnected to the input terminal Vi, and a lower-side Power MOSFET Q2 isconnected to a ground potential side. An LC smoothing filter, i.e., apower output filter consisting of an inductor L and a capacitor Co, anda CR smoothing filter consisting of a resistor R and a capacitor C areconnected in parallel to a midpoint of the Power MOSFETs Q1 and Q2.Moreover, the output terminal Vo is connected to a midpoint of the LCsmoothing filter, and one input (−) of an error amplifier EA isconnected to a midpoint of the CR smoothing filter. Here, the capacitorCo of the LC smoothing filter is a chip ceramic capacitor.

Also, a reference voltage Vref is connected to the other input (+) ofthe error amplifier EA. A pulse-width modulation (abbreviated as PWM)oscillator PWM, and gates of the Power MOSFETs Q1 and Q2 via a driverDRV are connected to an output of the error amplifier EA. The PowerMOSFETs Q1 and Q2 are driven in opposite phases to each other, and thusare electrically conducted alternately. In the present embodiment, anoutput voltage Vout is smaller than an input voltage Vin.

Next, the explanation will be given below regarding the circuitoperation in FIG. 1. The input voltage Vin applied to the input terminalVi is converted into a voltage by on/off controls over the upper-sidePower MOSFET Q1 and the lower-side Power MOSFET Q2 via the CR smoothingfilter. This converted voltage VFB is compared with the referencevoltage Vref by the error amplifier EA. As a consequence, an errorvoltage is generated in a state of being amplified at the output of theerror amplifier EA. This error voltage is converted into a PWM pulse bythe pulse-width modulation oscillator PWM. This PWM pulse is furtherconverted by the driver DRV into an on/off-time ratio (i.e., duty: α) atwhich the driver DRV drives the upper-side Power MOSFET Q1 and thelower-side Power MOSFET Q2. Moreover, a negative-feedback control isperformed over the PWM pulse so that the error voltage becomes equal to0. As a result of this, the converted voltage VFB becomes equal to thereference voltage Vref. In this case, the converted voltage VFB acquiredthrough the CR smoothing filter in the steady state is proportional tothe duty α of the input voltage Vin. Consequently, the followingrelational expression holds:

VFB=Vref=α·Vin

where the duty α assumes a value in the range of 0 to 1, since α isdefined as the on-time/(a total of the on-time and the off-time).

In the case of the ordinary step-down type converter, it has been foundout that the voltage-converted ratio in the steady state is equal to theratio, i.e., the duty, between the output voltage and the input voltage.Accordingly, assuming that the input voltage is Vin and the duty is α,the output from the LC smoothing filter, i.e., the output voltage Voutacquired at the output terminal Vo, can be determined by a relationalexpression:

Vout=α·Vin.

From the above-described 2 expressions, the following relationalexpression holds:

Vout=VFB=α·Vin.

Consequently, even if no direct negative-feedback control is performedover the output from the LC smoothing filter, if an indirect controlover the duty α using some other method proves successful, thissuccessful indirect control becomes equivalent to a direct control overthe output voltage Vout at the output terminal Vo. As a result, itbecomes possible to acquire, at the output terminal Vo, the voltage thatis proportional to the duty α of the input voltage Vin. In other words,the Power MOSFETs Q1 and Q2 are driven, thereby performing thenegative-feedback control over the output from the CR smoothing filter.This operation allows the desired voltage, which is proportional to theduty α of the input voltage Vin, to be also acquired at the output fromthe LC smoothing filter as the output voltage Vout.

As the voltage converting method based on the duty control over theupper-side Power MOSFET Q1 and the lower-side Power MOSFET Q2, thepresent embodiment is a primary-delay control method where the CRsmoothing filter is used for the control loop. Accordingly, since thereexists none of the secondary delay by the LC smoothing filter as wasfound in the prior art, the control loop does not become the oscillatingsystem. This prevents the oscillating waveform from occurring in theoutput, thereby making the loop stable. Consequently, according to thepresent embodiment, even if the chip ceramic capacitor with a small ESRis used as the capacitor of the LC smoothing filter, it is possible tostabilize the control loop.

Next, the explanation will be given below concerning the large-or-smallrelationship among the corner frequencies and the switching frequency ofthe above-described 2 smoothing filters. Let's assume that the cornerfrequency of the CR smoothing filter and that of the LC smoothing filterare equal to fCR and fLC respectively, and that the switching frequencyis equal to fSW. Then, setting the relationship among these frequenciesas fLC<fCR<fSW makes it possible to ensure the stability of the loop.Moreover, from this relationship, the feedback from the CR smoothingfilter results in a higher operation frequency as compared with thefeedback from the LC smoothing filter, which allows the implementationof the high-speed response. Also, fLC and fCR are set as frequenciesthat are different to some extent. This setting, even if the LCsmoothing filter's constants are modified, makes it unnecessary tochange the CR smoothing filter's constants, thereby allowing an increasein the degree-of-freedom of the design. With respect to the high-speedoperation of a 1-to-6-MHz switching frequency, values usable as the LCsmoothing filter's constants and the CR smoothing filter's constantsare, e.g., 0.2 μH, 220 μF, and 20 μF, 200 kΩ, respectively. If thevalues of these capacitors and this resistor are of these orders, itbecomes possible to mount (i.e., on-chip) the CR smoothing filter on asemiconductor integrated circuit chip, thereby makingexternally-attached components unnecessary. This means the following: Bymerely replacing the power-supply device illustrated in FIG. 1 by an ICwhose terminal location is the same (i.e., pin-compatible) as that ofthe power-supply control IC in the prior art, the printed wiring boardin the prior art can be utilized with no modification added thereto.

FIG. 2 is an explanatory diagram for explaining the chip layout in thecase where, in the power-supply device in FIG. 1, the CR smoothingfilter is built in a semiconductor chip. In FIG. 2, reference notationsC and R denote a built-in capacitor and a built-in resistor,respectively. These components are mounted on a semiconductor board thatis the same as the one that mounts thereon the error amplifier EA, thepulse-width modulation oscillator PWM, the driver DRV, and the PowerMOSFETs Q1 and Q2.

So far, the explanation has been given selecting, as the example, the CRsmoothing filter whose output is fed back to the error amplifier in thecontrol loop. Instead of the CR smoothing filter, however, the use ofanother high-response filter circuit allows the acquisition of basicallythe same effects. Also, although the explanation has been givenselecting the Power MOSFETs as the example of the semiconductorswitching components, the IGBTs may be used instead.

Embodiment 2

FIG. 3 illustrates the present embodiment. In FIG. 3, the same referencenotations are attached to the same configuration components in FIG. 1.The point in which FIG. 3 differs from FIG. 1 is that the CR smoothingfilter is set up at both ends of the inductor L of the LC smoothingfilter. In the present embodiment, since the electrostatic capacitanceof the capacitor Co of the output LC smoothing filter is large, theinductor-connected edge side of the capacitor Co can also be regarded asthe ground potential. The present embodiment also allows the acquisitionof basically the same effects in FIG. 1. Furthermore, the presentembodiment makes it possible to perform the negative feedback of aninfinitesimal capacitance change caused by a temperature change in thecapacitor Co of the LC smoothing filter. Consequently, even if the chipceramic capacitor with a small ESR is used, the present embodimentpermits an enhancement in the stability of the control loop. In thiscase as well, the constants of the embodiment in FIG. 1 are usable asthe CR smoothing filter's constants. FIG. 4 illustrates an explanatorydiagram for explaining the chip layout in the case where, in thepower-supply device in FIG. 3, the CR smoothing filter is built in asemiconductor chip.

Embodiment 3

FIG. 5 illustrates a power-supply device obtained by further providing atransient variation detecting circuit TVD into the 1st embodiment. Thistransient variation detecting circuit TVD controls the duty of thepulse-width modulation oscillator PWM by detecting a transient loadvariation between the output voltage Vout at the output terminal Vo anda voltage that results from adding a upper and lower limit-voltage width±Δ to the reference voltage Vref. FIG. 6 illustrates a concrete exampleof the pulse-width modulation oscillator PWM and that of the transientvariation detecting circuit TVD.

In FIG. 6, the pulse-width modulation oscillator PWM is a variableoscillator including a voltage-to-current converting circuit V/I,current-source MOSs 110, 120, inverters INV11, INV12, a capacitor 105,and a flip-flop FF. Also, the transient variation detecting circuit TVDincludes comparators CMP1, CMP2, switching MOSs SW1 to SW4, constantcurrent-sources I1 to I4, and inverters INV1 to INV8.

The transient variation detecting circuit TVD includes a wind comparatorconsisting of the 2 comparators CMP1, CMP2. The circuit TVD compares theoutput voltage Vout with the voltage that results from adding the upperand lower limit-voltage width ±Δ to the reference voltage Vref, therebydetecting the operation state of the output voltage Vout and determiningthe pulse duty α of the oscillator PWM indicated in FIG. 7. This meansthat, in the transient variation detecting circuit TVD, the controlmethod in the steady state and the one at the transient state areswitched into control modes that match the operation state.

From the outputs from the 2 comparators CMP1, CMP2, the following 3-wayinformation is acquired: (a) a case where the load current is decreased,(b) the steady state, (c) a case where the load current is increased.Using FIG. 7, these cases will be explained below:

The case (a) is under a condition Vout≧(Vref+Δ). At this time, theoutput duty α of the pulse-width modulation oscillator PWM is forcefullyset at 0%. For this purpose, the switching MOSs SW1 and SW4 are turnedon, and the switching MOSs SW3 and SW2 are turned off. As a result, acurrent from the constant current-source I1 is added to a current fromthe current-source MOS 110, then flowing together to the inverter INV11.A current from the constant current-source I4 is subtracted by a currentto the current-source MOS 120, so that the current value flowing to theinverter INV12 becomes equal to 0. Consequently, the upper-side PowerMOSFET Q1 is switched off, and the lower-side Power MOSFET Q2 isswitched on, which, eventually, makes the duty α equal to 0%. In thiscase, in order to set the duty α at 0% completely, it is preferable thatcurrent values from the constant current-sources I1 to I4 be each set atthe total current of differential pair operation currents of thevoltage-to-current converting circuit V/I.

The case (b) is under a condition (Vref+Δ)>Vout>(Vref−Δ). In this case,all of the switching MOSs SW1 to SW4 are turned off, and are operated inaccordance with a current ratio determined by a control instruction fromthe error amplifier EA. Since this current ratio is equal to the rate ofthe duty, the voltage that is proportional to the duty a of the inputvoltage Vin can be acquired as the output voltage Vout.

The case (c) is under a condition Vout≦(Vref−Δ), where the duty α isforcefully set at 100%. In this case, the switching MOSs SW3 and SW2 areturned on, and the switching MOSs SW1 and SW4 are turned off. As aresult, a current from the constant current-source I3 is added to thecurrent from the current-source MOS 120, then flowing together to theinverter INV12. A current from the constant current-source I2 issubtracted by the current to the current-source MOS 110, so that thecurrent value flowing to the inverter INV11 becomes equal to 0.Consequently, the upper-side Power MOSFET Q1 is switched on, and thelower-side Power MOSFET Q2 is switched off, which, eventually, makes theduty α equal to 100%. In this case, in order to set the duty α at 100%completely, it is preferable that the current values from the constantcurrent-sources I1 to I4 be each set at the total current of thedifferential pair operation currents of the voltage-to-currentconverting circuit V/I.

The present embodiment forcefully switches the duty α of the pulse-widthmodulation oscillator PWM to either 0% or 100% so that the voltagegenerated at the output terminal Vo at the transient state will fallwithin the upper and lower limit-voltage width ±Δ added to the referencevoltage Vref. This rapidly suppresses the output voltage Vout within(Vref±Δ). Moreover, when the operation state enters the steady state,the present embodiment causes the output voltage to be stabilized as thevoltage that is proportional to the duty α of the input voltage.

In this way, in the present embodiment, the control mode isautomatically switched depending on whether the operation state is thetransient state or the steady state. As a consequence, with respect toeven, e.g., an about 10A transient load variation having the highcurrent slew rate (i.e., di/dt) of 500A/μs, it becomes possible tosimultaneously implement both the high-speed response and thestabilization of the output voltage in the steady state.

Next, using FIG. 20, the description will be given below concerninganother embodiment of the pulse-width modulation oscillator PWM. Acircuit illustrated in FIG. 20 can be implemented by a combination of anoscillator OSC, a one-shot multivibrator OSM, and a V/I converter V/I. Aconstant time-period pulse can be generated by the oscillator OSC asfollows: A MOS 130 and a constant current-source I5 set a constantcurrent which is needed for determining the desired time-period. Next,this constant current is made to flow to the current-source MOSs 110,120 of the pulse-width modulation oscillator PWM in FIG. 6. Also, whenthis constant time-period pulse is applied to a clock terminal CLK ofthe one-shot multivibrator OSM, the terminal voltage of a capacitor CTbecomes equal to 0 on a temporary basis. At the next moment, however,the capacitor CT is electrically charged by a current that results fromconverting the error voltage of the error amplifier EA by the V/Iconverter V/I. Moreover, a time that has elapsed until this chargevoltage attains to a predetermined threshold value is acquired as thePWM pulse. In this way, the series of pulse-width modulation oscillatingoperations can be repeated. Namely, it becomes possible to acquire thePWM pulse that is proportional to the error voltage of the erroramplifier EA.

This pulse-width modulation oscillator PWM is used as an effective unitin a multi-phase control in FIG. 11 and FIG. 12 which will be describedlater. In this case, in order to implement the multi-phase operation, aphase shift circuit needs to be inserted after the oscillator OSC.

Embodiment 4

FIGS. 8 to 10 illustrate the present embodiment. The embodiment in FIG.8, which is obtained by providing the transient variation detectingcircuit TVD into the embodiment in FIG. 3, allows the acquisition ofbasically the same effects in FIG. 5. The configurations in FIG. 9 andFIG. 10 are as follows: In the circuit diagrams in FIG. 1 and FIG. 3,the input into the transient variation detecting circuit TVD is drawnfrom the midpoint of a series circuit that consists of a capacitor C3and a resistor R3 which are set up at both ends of the inductor L of theLC smoothing filter. As a result of this, the phase of an inductor Lcurrent, which can be detected by the series circuit of the capacitor C3and the resistor R3, and the charge/discharge phase of the outputcapacitor Co can be made to coincide with each other. Consequently, itbecomes possible to eliminate as much as possible excessive/redundantelectric charges produced by the charge/discharge of the outputcapacitor Co from the inductor L current. This makes it possible notonly to implement the high-speed response and the high stability, butalso to reduce a variation (i.e., ripple) in the output voltage at thetransient state.

Embodiment 5

The present embodiment is a multi-phase embodiment where the pluralpower-supply device units in the 1st to the 4th embodiments are operatedin parallel. The present embodiment combines the 2 or more same-typepower-supply devices indicated in the 1st to the 4th embodiments.Hereinafter, the explanation will be given below selecting the 2-phasingas the example.

FIG. 11 illustrates the embodiment that results from multi-phasing thepower-supply device unit in FIG. 8. In order to implement themulti-phasing, the embodiment in FIG. 11 newly includes the oscillatorOSC and a phase shift circuit PSFT, which generate 2-phase pulses whosephases are shifted to each other by 180°. This embodiment inputs each ofthe 2-phase pulses into each of pulse-width modulation oscillators PWM1and PWM2, thereby implementing the multi-phase control.

FIG. 12 illustrates, in more detail, the embodiment of the power-supplydevice in FIG. 11. In FIG. 12, the pulse-width modulation oscillatorPWM1 includes a voltage-to-current converting circuit V/I1 and aone-shot multivibrator OSM1. In the steady state, the oscillator PWM1operates by receiving a pulse signal from the phase shift circuit PSFT.

Using an operation state mode in FIG. 13, the explanation will be givenbelow regarding the operation of the embodiment in FIG. 12. Thisoperation state mode will be explained in much the same way as the caseof the 3rd embodiment. Hereinafter, the explanation will be givenconcerning the Phase 1 power-supply illustrated on the upper-half sidein FIG. 12.

(a) In the case of Vout≧(Vref+Δ), the output duty of the pulse-widthmodulation oscillator PWM1 is forcefully set at 0%. For this purpose,the reset RST of the one-shot multivibrator OSM1 is turned on, whichmakes the duty equal to 0%.

(b) In the case of (Vref+Δ)>Vout>(Vref−Δ), as an ordinary operation ofthe one-shot multivibrator, the OSM1 receives the pulse from the phaseshift circuit PSFT as a clock CLK, thereby generating an on-pulse width.The on-pulse width is determined by the current value from thecurrent-source MOS 210 and the capacitance value of a capacitor CT1,i.e., a timing capacitor. This on-pulse width is of a control mode thatoperates in accordance with the current ratio determined by the controlfrom the error amplifier EA. Namely, since this current ratio is equalto the duty, the output voltage Vout becomes equal to the voltage thatis proportional to the duty α of the input voltage Vin.

(c) In the case of Vout≦(Vref−Δ), the duty is forcefully set at 100%.For this purpose, both ends of the capacitor CT1, i.e., the timingcapacitor, are short-circuited by a MOS switch M21 so as to maintain theon-state, which makes the duty equal to 100%. Incidentally, a detectionresult by an overcurrent detecting circuit OC1 is also added to thereset RST, thereby preventing a component breakdown caused by anovercurrent from the upper-side Power MOSFET Q1. Concerning the Phase 2power-supply on the lower-half side in FIG. 12, the explanation will beomitted because the operation is the same as the Phase 1 power-supply.

In the operations described so far, in the steady state, the inductorcurrents from the 2 power-sources operate in opposite phases, i.e., inphases shifted to each other by 180°. Meanwhile, at the transient time,the inductor currents from the 2 power-supplies become the same in theirphases, thereby dealing with a rapid load variation. The presentembodiment not only increases the output current by using the pluralpower-supplies, but also reduces a ripple in the output voltage.

In the case of providing the 2 or more power-supply device units, thereare provided an oscillator and a phase shift circuit that the pluralpower-supply device units have in common. Moreover, in the steady state,phases of driving pulses of the upper-side/lower-side Power MOSFETs inthe respective power-supply device units are respectively shifted tophases that result from dividing 360° by the number of the power-supplydevice units located in parallel. At the transient state, as are thecases with the above-described (a) and (c), all of the parallelpower-supply device units are operated by driving pulses of one and thesame phase. In the case of, e.g., the 4 power-supply device units, it isadvisable to shift the phases to the respective phases of 0° (i.e.,criterion), 90°, 180°, and 270°.

Embodiment 6

Next, the explanation will be given below concerning an embodiment ofthe IC chip configuration of the power-supply control device in thepresent invention.

FIG. 14 illustrates the embodiment of the one-chip configuration of thecircuit configuration illustrated in FIG. 8. In FIG. 14, circuits andfunctions are all implemented on-chip on one semiconductor board exceptfor the following externally-mounted components: The LC smoothingfilter, the CR circuit consisting of the capacitor C3 and the resistorR3 for detecting the current phase of the transient variation detectingcircuit TVD, and a boost circuit consisting of a diode DBT and acapacitor CBT.

The on-chip implemented circuits and functions are as follows: The CRsmoothing filter consisting of the capacitor C and the resistor R, theerror amplifier EA, the reference voltage Vref, the pulse-widthmodulation oscillator PWM, a dead band circuit DBU, a dead band circuitDBL, a level shift circuit LS, a driver DRVU, a driver DRVL, theupper-side/lower-side Power MOSFETs Q1, Q2, an overcurrent detectingcircuit OC, the transient variation detecting circuit TVD, an upper andlower limit-voltage generating circuit VΔ, a soft-start circuit SS, anunder-voltage lockout circuit UVLO, and a power-good circuit PWRGD.Incidentally, instead of acquiring the reference voltage Vref from aband-gap reference circuit, the reference voltage Vref may be acquiredby receiving a digital signal corresponding to a VID (: VoltageIdentification) code, using an on-chip D/A converter illustrated in FIG.15. Although there exist not-illustrated circuits and functions, the1-chip power-supply control IC in the present embodiment is equippedwith the functions implemented in compliance with the VRM 9.1 expoundedby the Intel Corporation.

Although, in FIG. 14, the explanation has been given selecting the casewhere the upper-side Power MOSFET Q1 is the NMOS, the MOSFET Q1 may alsobe a PMOS. In this case, the externally-mounted boost circuit becomesunnecessary. However, since it is necessary to drive the gate of thePMOS at the electric potential from the input terminal Vi, avoltage-generating supply for this necessity is implemented on-chip.

The voltage fed to the input terminal Vi and the one fed to apower-supply terminal Vcc may be made equal to each other, e.g., 5V or12V. Otherwise, the voltages may be made different, e.g., 12V is fed tothe input terminal Vi, and 5V is fed to the power-supply terminal Vcc.When the voltage fed to the input terminal Vi and the one fed to thepower-supply terminal Vcc are different, 5V to the power-supply terminalVcc may be fed from the outside. Otherwise, 5V may be generated by theon-chip circuit from 12V fed to the input terminal Vi, then beingsupplied thereto. Incidentally, when feeding 12V to the input terminalVi, an about 7V Zener diode is connected to the boost circuit in FIG. 14in series with the diode DBT, thereby preventing the gate voltage of theupper-side Power MOSFET from becoming too large.

Also, in the operation of the soft-start circuit, at the time ofinjecting the power-supply, it is preferable to mask the output signalfrom the transient variation detecting circuit for the high-speedresponse.

Embodiment 7

FIG. 16 illustrates a multi-phase-compatible IC chip configuration inthe present embodiment. The configuration in FIG. 16 results frommulti-phasing the circuit configuration of the IC chip illustrated inFIG. 14. The point that differs from the 6th embodiment in FIG. 14 isthat the oscillator OSC and the phase shift circuit PSFT are added tothe IC chip. As IC pins that become necessary for implementing themulti-phasing, there exist terminals for providing its-own/the other ICchips with phase pulses φ1 to φ4 that correspond to the number of themulti phases, and terminals for supplying the reference voltage Vref,and outputs from the upper and lower limit-voltage generating circuit VΔto the transient variation detecting circuit TVD.

In the case of configuring the multi phases, at first, IC chips areprepared by the number of the desired multi phases, and, from among theIC chips, one IC chip is selected as a master. Concretely, a selectionsignal SEL0 for selecting the master IC chip activates the oscillatorOSC and a switch SWr, and 2 bits of selection signals SEL1 and SEL2specify the desired multi-phase number. Next, the master IC chipsupplies the phase pulses φ2 to φ4, the reference voltage Vref, and theoutputs Vref+Δ and Vref−Δ from the upper and lower limit-voltagegenerating circuit VΔ. As a result, it turns out that φ2 to φ4, Vref,Vref+Δ, and Vref−Δ are added to the other IC chips, respectively. Thisallows the implementation of the multi-phasing.

Although, in the present embodiment, the multi-phase number has beenillustrated as 4, no limitation is imposed on the multi-phase number.The selection-signal number for setting the multi-phase number ismodified, and the circuit configuration of the phase shift circuit PSFTis modified to a circuit configuration that matches the multi-phasenumber, and these pieces of information are installed into the IC chips.This allows the multi-phase number to be increased or decreaseddepending on the requirements.

Embodiment 8

FIG. 17 illustrates an embodiment where the power-supply control IC chipin the present invention is implemented on a printed wiring board. InFIG. 17, the power-supply control ICs, and the inductor L and thecapacitor Co are mounted on a printed wiring board PB with the use of aBGA (: Ball Grid Array) and chip components, respectively, therebyallowing the downsized high-density implementation. Here, the capacitorCo is the chip ceramic capacitor. Incidentally, although notillustrated, in addition to these components, the CR circuit of thecapacitor C3 and the resistor R3, the boost circuit, and the inputcapacitor are mounted on the printed wiring board PB with the use ofchip components in this embodiment. Also, other than the on-chipmounting by the BGA, the CSP (: Chip Size Package) mounting may also beemployed.

Furthermore, in the case of the multi-phase compatibility, other thanthe on-chip mounting of the plural power-supply control ICs, the MCM (:Multi Chip Module) mounting may also be employed. In addition to thesemountings, components divided onto 2 IC chips, such as a control unitincluding the error amplifier, the oscillator PWM, and the like, and adriver unit where the Power MOSFETs are built-in, may also be mounted onthe printed wiring board in much the same way.

As described above, according to the present embodiment, it becomespossible to implement the elimination of a pin neck, an enhancement inthe heat-dissipating capability, and the downsizing of the printedwiring board of the power-supply device.

Embodiment 9

FIG. 18 illustrates the present embodiment. FIG. 18 illustrates theembodiment that results from applying the present invention to HDDs (:Hard Disk Drives). Each of the HDDs includes a magnetic storage disk, amagnetic head, a magnetic-disk rotating drive, a magnetic-head drive, amagnetic-head position controller, and an input/output signalcontroller. DC—DC converters DC-DC1 to DC-DCn, i.e., the power-supplydevices described in the first to the eigth embodiments, supply electricpower to these HDDs HDD1 to HDDn. As the DC—DC converters DC-DC1 toDC-DCn, i.e., the power-supply devices illustrated in FIG. 18, thesingle-phase power-supply devices or the multi-phase power-supplydevices are used, depending on the current capacities of the HDDS, i.e.the targets of the power supply.

Embodiment 10

Next, the explanation will be given below concerning an embodiment wherethe control scheme in the present invention is applied to isolation typeDC—DC converters. FIG. 19 illustrates the embodiment applied to aforward type converter. In FIG. 19, as is the case with FIG. 3, the CRsmoothing filter of C and R is set up at both ends of an inductor L ofthe forward type converter. Next, the error amplifier EA generates anamplified error voltage, using the relationship between a voltage VFB atthe midpoint of the CR smoothing filter and a reference voltage Vref.Moreover, the use of the pulse-width modulation oscillator PWM convertsthis amplified error voltage into a PWM pulse. This PWM pulse is passedthrough a transformer T2, and is applied to the gate of a Power MOSFETQD for driving a transformer T1, then being subjected to thenegative-feedback control. This allows a desired output voltage to beacquired in a stationary manner at the output terminal Vo. The presentmethod performs no negative-feedback control over the output from thepower LC filter, thereby making it possible to configure a highloop-stability power-supply system. Consequently, the present embodimentis especially effective when the ceramic capacitor is used as C of theLC filter.

Although, so far, the explanation has been given using the CR filter inFIG. 3, the explanation is also possible using the method in FIG. 1.Also, instead of the transformer T2, the implementation is also possibleusing a photo coupler. In FIG. 19, the explanation has been givenselecting the 1-stone forward type converter. The above-describedcontrol scheme, however, is also applicable to the other isolation typeDC—DC converters such as 2-stone forward type, push-pull type,half-bridge type, and full-bridge type.

Embodiment 11

Next, the illustration will be given below regarding an embodiment wherethe control scheme in the present invention is applied to acommercially-available power-supply IC. FIG. 21 illustrates the casewhere a PWM control IC HIP6311A and a driver-built-in Power MOSFET ICILS6571 of the Intersil Corporation are used as thecommercially-available power-supply IC. The midpoint of C and R ofone-side CR smoothing filter set up at both ends of an inductor L isconnected to a feedback terminal FB of the PWM control IC HIP6311A. Themidpoint of C3 and R3 of the other-side CR smoothing filter is connectedto a transient variation detecting circuit TVD comprised of a referencepower-source LT1790A and a converter LT1715 of the Linear TechnologyCorporation through a high-input-impedance buffer amplifier BA and aresistor RN. Moreover, from the relationship between logical levels “H”and “L” of two signals a and b acquired by the transient variationdetecting circuit TVD, 3 operation state modes, i.e., a PWM pulse signalPWM1 (desired duty α) outputted from the PWM control IC, a duty 0% α0,and a duty 100% α100, are switched selectively as indicated in FIG. 22by a selector HD74HC153, HD74HC157. Furthermore, its selected signal Yis outputted to a PWM terminal of the driver-built-in Power MOSFET IC.This shows that the control scheme in the present invention is alsoapplicable easily to the power-supply device configured using thecommercially-available power-supply IC. The application of the presentinvention is not limited to the products described in theabove-described embodiment. Incidentally, when the transient variationdetecting circuit TVD is not used, the PWM pulse signal PWM1 outputtedfrom the PWM control IC is directly connected to the PWM terminal of thedriver-built-in Power MOSFET IC, thereby making it possible to implementthe present invention.

It is needless to say that, although not illustrated, the power-supplydevices in the first to the eighth embodiments can be applied andexpanded to the other apparatuses, e.g., a VRM, a DC—DC converter forportable appliances, and a general-purpose DC—DC converter.

In the power-supply device of the present invention, none of thesecondary delay by the power LC smoothing filter enters the controlloop, which enhances the stability of the control loop. This furthermakes it possible to use the small-ESR chip ceramic capacitor in the LCsmoothing filter, thereby implementing the downsizing of thepower-supply device.

In the power-supply device of the present invention, the upper and lowerlimit value detecting circuit controls the high-speed response at thetransient state. This allows the power-supply device to make response toeven the high current slew rate (i.e., di/dt).

The power-supply device of the present invention can be easilymulti-phased. This makes it possible to simultaneously implement boththe large output current and the ripple-voltage reduction.

It should be further understood by those skilled in the art thatalthough the foregoing description has been made on embodiments of theinvention, the invention is not limited thereto and various changes andmodifications may be made without departing from the spirit of theinvention and the scope of the appended claims.

What is claimed is:
 1. A power-supply device including a step-down DC—DCconverter, comprising: power semiconductor switching components, drivingmeans for driving said power semiconductor switching components, apulse-width modulation oscillator for supplying said driving means witha driving signal, and an error amplifier for supplying said pulse-widthmodulation oscillator with an error signal indicating a comparisonresult between a reference value and an output power, wherein a controlloop of said power-supply device includes a filter providedindependently of a power output filter through which said output powerpasses, an output signal corresponding to said output power being fedback to said error amplifier alter having passed through saidindependently provided filter.
 2. The power-supply device according toclaim 1, wherein said power output filter is an LC filter consisting ofan inductor and a capacitor, said independently provided filter being aCR filter consisting of a capacitor and a resistor, said CR. filterbeing connected to said LC filter in parallel, and said output signalbeing fed back to said error amplifier after having passed through saidCR. filter.
 3. The power-supply device according to claim 1, whereinsaid power output filter is an LC filter consisting of an inductor and acapacitor, said independently provided filter being a CR filterconsisting of a capacitor and a resistor, said CR filter being connectedto both ends of said inductor of said power output filter, and saidoutput signal being fed back to said error amplifier after having passedthrough said CR filter.
 4. The power-supply device according to claim 2further comprising a transient variation detecting circuit, saidtransient variation detecting circuit detecting said output power froman output terminal of said power output filter, and, if said outputpower has been found to exceed a predetermined upper-limit value,outputting a signal for setting the duty α of said pulse-widthmodulation oscillator at 0%, and, if said output power has been found tobe lower than a predetermined lower-limit value, outputting a signal forsetting said duty a of said pulse-width modulation oscillator at 100%.5. The power-supply device according to claim 2, further comprising atransient variation detecting circuit, said transient variationdetecting circuit detecting said output power from an output terminal ofsaid CR filter newly provided at both ends of said inductor of saidpower output filter, and, it said output power has been found to exceeda predetermined upper-limit value, outputting a signal for setting theduty α of said pulse-width modulation oscillator at 0%, and, if saidoutput power has been found to be lower than a predetermined lower-limitvalue, outputting a signal for setting said duty α of said pulse-widthmodulation oscillator at 100%.
 6. The power-supply device according toclaim 1, wherein said power semiconductor switching components, saiddriving means for driving said power semiconductor switching components,said pulse-width modulation oscillator, said error amplifier, and atransient variation detecting circuit are formed on one and the samesemiconductor board, said transient variation detecting circuitdetecting said output power from an output terminal of said power outputfilter. and, if said output power has been found to exceed apredetermined upper-limit value, outputting a signal for setting theduty α of said pulse-width modulation oscillator at 0%. and, if saidoutput power has been found to be lower man a predetermined lower-limitvalue, outputting a signal for setting said duty α of said pulse-widthmodulation oscillator at 100%.
 7. A power-supply device implemented byapplying said power-supply device according to claim 1 to an isolationtype DC—DC converter.
 8. A power-supply device implemented by using acombination of an oscillator, a one-shot multivibrator, and avoltage-to-current converter as said pulse-width modulation oscillatorof said power-supply device according to claim
 1. 9. A multi-phasecontrol power-supply device, comprising: a phase shift circuit, and saidpulse-width modulation oscillator of said power-supply device accordingto claim 8, wherein said phase shift circuit is provided after saidoscillator, said one-shot multivibrator being provided in correspondencewith each phase.
 10. A power-supply device according to claim 1, whereinsaid error amplifier has a low input impedance, and is connected to theoutput of said independently provided filter through a buffer amplifierhaving a high input impedance.
 11. An integrated circuit wherein apower-supply device according to claim 1 is built in a semiconductorchip.
 12. The power-supply device according to claim 3, wherein arelation of fLC<fCR is established where the frequency of said CR filteris fCR and the frequency f said filter LC is fLC.
 13. A power-supplydevice of a step-down type DC—DC converter including power-supply deviceunits in plural number, each of said plural power-supply device units,comprising: power semiconductor switching components, driving means fordriving said power semiconductor switching components, a pulse-widthmodulation oscillator for supplying said driving means with a drivingsignal, and an error amplifier for supplying said oscillator with anerror signal of an output power, wherein each of said pluralpower-supply device units includes a filter provided independently of apower output filter through which said output power passes, an outputsignal being fed back to said error amplifier after having passedthrough said independently provided filter.
 14. The power-supply deviceaccording to claim 13, wherein, in order to perform a parallel operationof said plural power-supply device units, said plural power-supplydevice units include said pulse-width modulation oscillators in commontherewith, phases of said output driving signals from said pulse-widthmodulation oscillators being shifted, and said signals of which phaseshave been shifted being supplied to said plural power-supply deviceunits.
 15. The power-supply device according to claim 14, wherein eachof said plural power-supply device units further comprises a transientvariation detecting circuit, said transient variation detecting circuitdetecting said output power from an output terminal of said power outputfilter, and, if said output power has been found to exceed apredetermined upper-limit value, outputting a signal for setting theduty a of said pulse-width modulation oscillator at 0%, and, if saidoutput power has been found to be lower than a predetermined lower-limitvalue, outputting a signal for setting said duty a of said pulse-widthmodulation oscillator at 100%.
 16. A hard-disk device which includes amagnetic storage disk, a magnetic head, a magnetic-disk rotating drive,a magnetic-head drive, a magnetic-head position controller, aninput/output signal controller, and a power-supply device for supplyingpower, said power-supply device being a step-down type DC—DC converterwhich comprises power semiconductor switching components, driving meansfor driving said power semiconductor switching components, a pulse-widthmodulation oscillator for supplying said driving means with a drivingsignal, and an error amplifier for supplying said oscillator with anerror signal of an output power, wherein said step-down type DC—DCconverter further comprises a power output filter through which saidoutput power passes, and a filter provided independently of said poweroutput filter, an output signal being fed back to said error amplifierafter having passed through said independently provided filter, andwherein said power output filter is an LC filter consisting of aninductor and a capacitor, said independently provided filter being a CRfilter consisting of a capacitor and a resistor, said CR filter beingconnected to said LC filter in parallel, and said output signal beingfed back to said error amplifier after having passed through said CRfilter such that the feedback to said error amplified is through said CRfilter but not through said LC filter.